LO generation and distribution in a multi-band transceiver

ABSTRACT

A VCO of a PLL outputs a first differential signal of frequency FVCO. A first divide-by-two circuit local to the VCO divides the first differential signal and outputs a first quadrature signal of frequency FVCO/2. Two of the component signals of the first quadrature signal are routed to a second divide-by-two circuit local to a first mixer of a first device. The second divide-by-two circuit outputs a second quadrature signal of frequency FVCO/4 to the first mixer. All four signals of the first quadrature signal of frequency FVCO/2 are routed through phase mismatch correction circuitry to a second mixer of a second device. In one example, FVCO is a tunable frequency of about ten gigahertz, the first device is an IEEE802.11b/g transmitter or receiver that transmits or receives in a first band, and the second device is an IEEE 802.11a transmitter or receiver that transmits or receives in a second band.

CLAIM OF PRIORITY

The present application is a continuation application of, and patentclaims priority to, U.S. Utility patent application Ser. No. 12/948,166entitled “LO Generation and Distribution in a Multi-Band Transceiver”filed, Nov. 17, 2010 the entire disclosure of which is hereby expresslyincorporated by reference and assigned to Qualcomm Inc.

BACKGROUND

1. Field

The present disclosure relates to the generation and distribution oflocal oscillator signals within a multi-band transceiver.

2. Background

Cellular telephone handsets are sometimes made to have a multi-band WiFicommunication capability where the WiFi communication capability is tobe compliant with multiple Institute of Electrical and ElectronicEngineers (IEEE) standards including IEEE802.11a, IEEE802.11b, andIEEE802.11g. Due to the application in a cellular telephone handset,reducing the integrated circuit area consumed by the multi-bandtransceiver is important to reduce cost. Maintaining low powerconsumption is also important to increase talk time. If the multi-bandtransceiver is to operate in compliance with the IEEE802.11b andIEEE802.11g standards, then it should be able to receive and to transmitsignals in the so-called 2.5 GHz band. This band actually extends from alower bound of approximately 2.412 GHz to an upper bound ofapproximately 2.484 GHz. If the multi-band transceiver is to operate incompliance with the IEEE802.11a standard, then it should be able toreceive and to transmit signals in the so-called 5.0 GHz band. This bandactually extends from a lower bound of approximately 4.915 GHz to anupper bound of approximately 5.825 GHz.

The upconversion and downconversion processes that occur in themulti-band WiFi transceiver generally require both I and Q localoscillator signals in the frequency of the band of interest, where the Ilocal oscillator signal is differential and where the Q local oscillatorsignal is differential. Accordingly, four phases (0, 90, 180, 270degrees) of a first tunable quadrature local oscillator signal around2.5 GHz are typically required for IEEE802.11b/g band operation, andfour phases (0, 90, 180, 270 degrees) of a second tunable quadraturelocal oscillator signal around 5.0 GHz are typically required forIEEE802.11a band operation. These tunable local oscillator signals aretypically generated using a Phase-Locked Loop (PLL) that in turnincludes a Voltage Controlled Oscillator (VCO). For cost reasons, thePLL and VCO are realized on the same integrated circuit as are the PowerAmplifiers (PAs) that output the high power signals to the transmitterantenna. Unfortunately, a strong transmitter output signal from a PA canbe injected back such that it disturbs the VCO if the VCO is operatingat the same approximate frequency as the PA output signal frequency.This disturbance of the VCO can be due to injection back into the VCOthrough the power supply conductors, through ground conductors, throughthe integrated circuit substrate, or due to inductive coupling between aPA coil and the coil of the VCO. To prevent such unwanted interactionbetween a PA output signal and the VCO, an architecture is typicallyemployed in which the VCO does not operate at the same frequency as thefrequency of the PA output signal. There are several architectures foraccomplishing this.

A first architecture involves running the VCO in the ten gigahertz rangeand routing the VCO output signal to the transmitters and receivers. Inthe case of 5.0 GHz band transmitters and receivers that requirequadrature local oscillator signals at 5.0 GHz, a circuit close to thetransmitter or receiver receives the ten gigahertz signal and generatesthe 5.0 GHz quadrature local oscillator signal required. In the case oftransmitters and receivers that require quadrature local oscillatorsignals at 2.5 GHz, a circuit close to the transmitter or receiverdivides the ten gigahertz signal by four and generates the 2.5 GHzquadrature signal. This simple architecture is generally not usedbecause it has a very high power consumption due to parasitics in therouting and because it suffers from reliability and yield problems dueto the high frequency of operation in the LO distribution network.

FIG. 1 (Prior Art) is a diagram of a second architecture for generatingthe required local oscillator signals using a single local oscillatorwithout the oscillator being unduly adversely affected by the poweramplifier output signal. This architecture is sometimes referred to asthe “offset LO” architecture. A VCO 1 outputs a differential signal of afrequency that is ⅔ the frequency of the desired PA output signal. ThePA of an IEEE802.11a transmitter is identified by reference numeral 2.The VCO output signal is then divided down by two by a divider 3 togenerate quadrature signals at ⅓ of the desired PA output frequency. Apolyphase filter 4 is used to generate quadrature signals at ⅔ thedesired PA output frequency. A mixer 5 mixes the quadrature signals of ⅔the desired PA output frequency with the quadrature signals of ⅓ thedesired PA output frequency to generate a differential signal 6 of thedesired PA output frequency. A first polyphase filter 7 is used togenerate the quadrature signals that are supplied to the mixer 8 of anIEEE802.11a transmitter 9 portion of the circuit. A second polyphasefilter 10 is used to generate the quadrature signals that are suppliedto the mixer 11 of an IEEE802.11a receiver 12 portion of circuit. Bytuning the VCO output signal frequency in the range of fromapproximately 3.27 GHz to 3.88 GHz, the local oscillator signalssupplied to the mixers 8 and 11 of the IEEE802.11a transmitter and theIEEE802.11a receiver can be set to have a desired frequency in a tuningrange of from about 4.915 GHz to about 5.825 GHz as required forIEEE802.11a band operation.

To generate the quadrature local oscillator signals for IEEE802.11b/gband operation, an additional divide-by-two circuit 13 is provided.Divide-by-two circuit 13 generates quadrature signals at half thefrequency of the signal output by mixer 5. These quadrature signals areprovided to the mixer of an IEEE802.11b/g band transmitter (not shown),and are also provided to the mixer of an IEEE802.11b/g band receiver(not shown). The IEEE802.11b/g band transmitter can be considered tohave the same topology as the transmitter 9. The IEEE802.11b/g bandreceiver can be considered to have the same topology as the receiver 12.By tuning the VCO frequency in a tuning range from 3.618 GHz to 3.726GHz, the frequency of the local oscillator signals supplied to mixers ofthe IEEE802.11b/g band transmitter and the IEEE802.11b/g receiver can beset to have a desired frequency in a tuning range of from about 2.412GHz to about 2.484 GHz as required for IEEE802.11b/g band operation. The“offset LO” architecture of FIG. 1 is desirable in that the VCO operatesat a different frequency from the frequency of the transmitter outputsignal where this different frequency is not a multiple of the poweramplifier output signal. This offset LO architecture, however, hasdrawbacks in that it is expensive to implement. It also exhibits fairlyhigh current consumption. These two drawbacks make it undesirable foruse in a cellular handset application.

FIG. 2 (Prior Art) is a third architecture for generating the requiredlocal oscillator signals for multi-band WiFi operation using a singlelocal oscillator. This architecture is sometimes referred to as the“heterodyne LO” architecture. A VCO 14 outputs a signal at ⅔ thefrequency of the desired PA output signal as in the case of the offsetLO architecture, but in the case of the heterodyne LO architecture thereare two cascaded mixers 15 and 16 in the transmit signal path of theIEEE802.11a transmitter 17 and there are two cascaded mixers 18 and 19in the receive signal path of the IEEE802.11a receiver 20. In the caseof the transmitter, the first mixer 15 upconverts by mixing the basebandsignal to be transmitted with the quadrature signal 21 of ⅓ the desiredPA output signal frequency. The second mixer 16 then further upconvertsby mixing the output of the first mixer with the differential signal 22of ⅔ of the desired PA output signal frequency as output by the VCO. Theresult of the cascaded mixing is the same as if a single upconvertingmixer were used to mix the baseband transmit signal with a quadraturesignal of the desired PA output signal frequency. The inverse processoccurs in the IEEE802.11a receiver 20. Reference numeral 23 identifiesthe power amplifier of the IEEE802.11a transmitter 17.

To generate the quadrature local oscillator signals for IEEE802.11b/gband operation, an additional mixer 24 and a divide-by-two circuit 25are provided as illustrated. The quadrature signals from divide-by-twocircuit 25 are supplied to the mixer of a homodyne direct conversiontransmitter (not shown). This transmitter is used for IEEE802.11b/g bandtransmitting. Similarly, the quadrature signals from divide-by-twocircuit 25 are supplied to the mixer of a homodyne direct conversionreceiver (not shown). This receiver is used for IEEE802.11b/g bandreceiving. By tuning the VCO frequency in the tuning range from 3.618GHz to 2.726 GHz, the frequency of the local oscillator signals suppliedto mixers of the IEEE802.1b/g transmitter and receiver can be setappropriately for IEEE802.11b/g band operation. In the heterodyne LOarchitecture of FIG. 2, the active PA outputs its powerful output signalat a frequency that is different from the VCO operating frequency. TheVCO operates at a frequency that is not an even multiple of the PAoutput signal frequency and this reduces the unwanted influence of thePA output signal on the VCO. Unfortunately, the heterodyne LOarchitecture of FIG. 2 is also expensive to implement and has arelatively large current consumption. A further drawback is that signalquality may be compromised due to additional unwanted tones generated bythe additional mixing circuitry involved.

SUMMARY

Within the local oscillator of a multi-band radio transceiver, a VoltageControlled Oscillator (VCO) of a Phase-Locked Loop (PLL) outputs a firstdifferential signal of a tunable frequency FVCO. In one example, thistunable frequency FCVO is close to the upper frequency limit that a VCOcan be made to use and still have acceptable power consumption andreliability, given the semiconductor technology being used to implementthe transceiver.

A first divide-by-two circuit local to the VCO divides this firstdifferential signal of frequency FVCO by two and outputs a firstquadrature signal of frequency FVCO/2. The first quadrature signalinvolves four component signals. Two of these four component signals arerouted to a second divide-by-two circuit local to a first mixer of afirst device. The second divide-by-two circuit outputs a secondquadrature signal of frequency FVCO/4 to the first mixer. All fourcomponent signals of the first quadrature signal of frequency FVCO/2 arerouted from the first divide-by-two circuit and through a first phasemismatch correction circuit to a second mixer of a second device. In oneexample, FVCO is a tunable frequency of about ten gigahertz, the firstdevice is an IEEE802.11b/g transmitter or receiver that transmits orreceives in a first frequency band (2.412 GHz to 2.484 GHz), and thesecond device is an IEEE802.11a transmitter or receiver that transmitsor receives in a second frequency band (4.915 GHz to 5.825 GHz).

An on-chip internal test loop back connection is also provided. Thistest loop back connection allows the receiver on the chip for the secondfrequency band to receive and detect a signal output by the transmitteron the chip for the second frequency band. After the loop back signalhas been received in this way and has been demodulated, evidence ofphase mismatch in the components of the quadrature LO signals of thetransmitter and/or receiver is detected. The first phase mismatchcorrection circuits in the quadrature LO signal paths going to thetransmitter and/or receiver are then controlled so that the relativephases of the component signals are adjusted such that the detectedphase mismatch is reduced or eliminated.

In addition to the circuitry described above involving two devices, twoof the four component signals of the first quadrature signal are alsorouted to a third divide-by-two circuit local to a third mixer of athird device. The third divide-by-two circuit outputs a third quadraturesignal of frequency FVCO/4 to the third mixer. All four componentsignals of the first quadrature signal of frequency FVCO/2 are alsorouted from the first divide-by-two circuit and through a second phasemismatch correction circuit to a fourth mixer of a fourth device. In oneexample, the third device is an IEEE802.11b/g receiver that receives inthe first frequency band and the fourth device is an IEEE802.11areceiver that receives in the second frequency band. The VCO, the firstdivide-by-two circuit, the second divide-by-two circuit, the thirddivide-by-two circuit, the first phase mismatch correction circuit, thesecond phase mismatch correction circuit, the first device, the seconddevice, the third device and the fourth device are all disposed on thesame integrated circuit die.

Note that the lower 2.412 GHz bound of the first frequency band isapproximately half the frequency of the lower 4.915 GHz bound of thesecond frequency band, and note that the upper 2.484 GHz bound of thefirst frequency band is approximately half of the frequency of the upper5.825 GHz bound of the second frequency band. The novel LO generationand distribution architecture set forth here takes special advantage ofthis fact, taken together with the fact that twice the frequency of theupper bound of the higher frequency band is about ten gigahertz and thishigh frequency is the highest frequency that a suitable VCO can be madeto operate at with reasonable power consumption and reliability for thecellular telephone handset application. Although the VCO operates at atunable frequency of approximately ten gigahertz, high frequency tengigahertz signals are not routed over substantial distances across thedie from the VCO to the first, second, third or fourth devices. Where2.5 GHz quadrature signals are required, two component signals of the5.0 GHz quadrature signal are routed as a differential signal over thesubstantial distance and the needed 2.5 GHz quadrature signals aregenerated close to where the quadrature signals are needed using adivide-by-two circuit. Where 5.0 GHz quadrature signals are required,the four component signals of the 5 GHz quadrature signal are routedover the substantial distance to the circuit that needs the 5.0 GHzquadrature signal, but phase mismatch correction circuitry is used tocorrect phase mismatch problems that might arise due to communicatingthe 5.0 GHz quadrature signal that distance.

The coupling gain of signals into the tank of the VCO is made to dropoff rapidly for signals above ten gigahertz so that any harmonics ofPower Amplifier (PA) output signals above ten gigahertz will not undulyperturb VCO operation. Even order harmonics of the 5.0 GHz and 2.5 GHzPA output signals may land on the ten gigahertz operating frequency ofthe VCO, but they have little impact on VCO operation due to theirrelative low strengths (as compared to the signal strengths of odd orderharmonics) and due to the differential nature of the routing circuitry.As compared to offset LO architecture of the prior art and as comparedto the heterodyne LO architecture of the prior art, the novel LOgeneration and distribution circuitry described above can be realized ina comparatively small amount of integrated circuit area and can be madeto have a comparatively small current consumption.

The foregoing is a summary and thus contains, by necessity,simplifications, generalizations and omissions of detail; consequently,those skilled in the art will appreciate that the summary isillustrative only and does not purport to be limiting in any way. Otheraspects, inventive features, and advantages of the devices and/orprocesses described herein, as defined solely by the claims, will becomeapparent in the non-limiting detailed description set forth herein.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 (Prior Art) is a diagram of an offset LO architecture forgenerating local oscillator signals for a multi-band transceiver.

FIG. 2 (Prior Art) is a diagram of a heterodyne LO architecture forgenerating local oscillator signals for a multi-band transceiver.

FIG. 3 is a high-level diagram of a multi-band IEEE802.11 transceiverthat employs a divide-by-two LO generation and distribution architecturein accordance with one novel aspect.

FIG. 4 is a diagram of the RF transceiver integrated circuit of FIG. 3.

FIG. 5 is a diagram of the local oscillator within the RF transceiverintegrated circuit of FIG. 4.

FIG. 6 is a more detailed diagram of the LO generation and distributioncircuitry within the RF transceiver integrated circuit of FIG. 4.

FIG. 7 is a simplified diagram of the LO generation and distributioncircuitry within the RF transceiver integrated circuit of FIG. 4.

FIG. 8 is a diagram of one of the programmable delay lines in the LOgeneration and distribution circuitry of FIG. 6.

FIG. 9 is a high-level diagram of one of the divide-by-two circuits inthe LO generation and distribution circuitry of FIG. 6.

FIG. 10 is a more detailed diagram of the divide-by-two circuit of FIG.9.

FIG. 11 is a circuit diagram of one of the latches of the divide-by-twocircuit of FIG. 9.

FIG. 12 is a diagram that shows operation of the latch of FIG. 11 in thetracking mode.

FIG. 13 is a diagram that shows operation of the latch of FIG. 11 in thelocking mode.

FIG. 14 is a high-level diagram of one of the programmable drivers ofthe programmable delay lines in the LO generation and distributioncircuitry of FIG. 6.

FIG. 15 is a diagram of one of the drivers within the programmabledriver of FIG. 14.

FIG. 16 is a circuit diagram of the driver of FIG. 15.

FIG. 17 is a table that compares the amount of integrated circuit arearequired to implement the offset LO architecture of FIG. 1, theheterodyne LO architecture of FIG. 2, and the embodiment of FIG. 6.

FIG. 18 is a table that compares the current consumption of the offsetLO architecture of FIG. 1, the heterodyne LO architecture of FIG. 2, andthe embodiment of FIG. 6.

FIG. 19 is a flowchart of a method 500 in accordance with one novelaspect.

DETAILED DESCRIPTION

FIG. 3 is a diagram of a multi-band IEEE802.11 mobile station device(STA) 100 transceiver embodied in a cellular telephone handset. TheIEEE802.11 transceiver is called “multi-band” in that it can operate inaccordance with the IEEE802.11a standard using a frequency band from4.915 GHz to 5.825 GHz or it can operate in accordance with theIEEE802.11b/g standard using a frequency band from 2.412 GHz to 2.484GHz. For simplification purposes, the lower band from 2.412 GHz to 2.484GHz band is referred to hereinafter as the “first frequency band” andthe higher band from 4.915 GHz to 5.825 GHz is referred to hereinafteras the “second frequency band”. IEEE802.11b/g is referred to as the“first standard”. IEEE802.11a is referred to as the “second standard”.Transceiver 100 includes (among other parts not illustrated) an antenna101, an RF transceiver integrated circuit 102, and a digital basebandintegrated circuit 103. RF transceiver integrated circuit 102 is calleda “transceiver” because it includes both transmitter circuitry as wellas receiver circuitry.

FIG. 4 is a more detailed block diagram of the RF transceiver integratedcircuit 102 of FIG. 3. The transceiver includes a first transmitter 104for transmitting signals compliant with the first standard, a secondtransmitter 105 for transmitting signals compliant with the secondstandard, a first receiver 106 for receiving signals compliant with thefirst standard, and a second receiver 107 for transmitting signalscompliant with the second standard. Transmitter 104 and receiver 106 areused when the mobile station 100 is communicating using the firststandard, whereas transmitter 105 and receiver 107 are used when themobile station is communicating using the second standard. In additionto the transmitters and receivers, the RF transceiver integrated circuit102 includes serial bus interface circuitry 108, local oscillatorcircuitry 109, two power amplifiers 117 and 127, and twotransmit/receive switches 128 and 118. The local oscillator circuitry109 includes a Phase-Locked Loop (PLL) 110 that in turn includes aVoltage Controlled Oscillator (VCO) 111.

If mobile station 100 is transmitting a signal in accordance with thefirst standard, then information to be transmitted is converted intoanalog form by a Digital-to-Analog Converter (DAC) 112 (see FIG. 3) inthe digital baseband integrated circuit 103 and is supplied viaconductors 113 to transmitter 104. Baseband filter 114 filters out noisedue to the digital-to-analog conversion process. Mixer 115 under controlof local oscillator 109 then up-converts the signal into a highfrequency signal. Driver amplifier 116 and power amplifier 117 amplifythe high frequency signal. The resulting signal passes through T/Rswitch 118 and diplexer 119 to antenna 101 so that a high frequency RFsignal 120 is transmitted from antenna 101. The control signals thatcontrol the T/R switches 118 and 128 are omitted from the diagram.Driver amplifier 116 in this case is a power amplifier from theperspective of outputting a signal of such power that preventing theoutput signal from perturbing VCO operation in the LO is of importance.The digital baseband integrated circuit 103 controls the transmitter 104by controlling the frequency of a local oscillator quadrature signal LO1121 supplied to mixer 115. LO1 in this case includes a differential Isignal involving signals IP and IN, as well as a differential Q signalinvolving signals QP and QN.

If mobile station 100 is transmitting a signal in accordance with thesecond standard, then information to be transmitted is converted intoanalog form by DAC 122 in the digital baseband integrated circuit 103and is supplied via conductors 123 to transmitter 105. Baseband filter124 filters out noise due to the digital-to-analog conversion process.Mixer 125 under control of local oscillator 109 then up-converts thesignal into a high frequency signal. Driver amplifier 126 and poweramplifier 127 amplify the high frequency signal. The high frequencysignal passes through T/R switch 128 and diplexer 119 to antenna 101 sothat a high frequency RF signal 129 is transmitted from antenna 101.Driver amplifier 126 is a power amplifier. The digital basebandintegrated circuit 103 controls the transmitter 105 by controlling thefrequency of a local oscillator signal LO2 130 supplied to mixer 125.

When mobile station 100 is receiving a signal in accordance with thefirst standard, a high frequency RF signal 131 is received on antenna101. Information from signal 131 passes through diplexer 119, T/R switch118, matching network 132, and through the receiver 106. The signal isamplified by Low Noise Amplifier (LNA) 133 and is down-converted infrequency by mixer 134. The resulting down-converted signal is filteredby baseband filter 135 and is passed to the digital baseband integratedcircuit 103 via conductors 136. An Analog-to-Digital Converter (ADC) 137in the digital baseband integrated circuit 103 converts the signal intodigital form and the resulting digital information is processed bydigital circuitry in the digital baseband integrated circuit 103. Thedigital baseband integrated circuit 103 tunes the receiver 106 bycontrolling the frequency of the Local Oscillator signal (LO3) 138supplied to mixer 134.

If mobile station 100 is receiving a signal in accordance with thesecond standard, then a high frequency RF signal 139 is received onantenna 101. Information from signal 139 passes through diplexer 119,T/R switch 128, matching network 140, and through the receiver 107. Thesignal is amplified by LNA 141 and is down-converted in frequency bymixer 142. The resulting down-converted signal is filtered by basebandfilter 143 and is passed to the digital baseband integrated circuit 103via conductors 144. An ADC 145 in the digital baseband integratedcircuit 103 converts the signal into digital form and the resultingdigital information is processed by digital circuitry in the digitalbaseband integrated circuit 103. The digital baseband integrated circuit103 tunes the receiver 107 by controlling the frequency of the LocalOscillator signal (LO4) 146 supplied to mixer 142.

A processor 151 in digital baseband integrated circuit 103 controlslocal oscillator 109 and the frequencies of LO signals 121, 130, 138 and146 by sending appropriate control information to the RF transceiverintegrated circuit 102 via bus mechanism 152, serial bus interface 147,across a serial bus 148, through serial bus interface 108, and controllines 150. Processor 151 accesses and executes program 230 ofprocessor-executable instructions stored in semiconductor memory 231.Semiconductor memory 231 is a processor-readable medium that isaccessible by processor 151 via bus mechanism 152.

FIG. 5 is a more detailed diagram of local oscillator 109. APhase-Frequency Detector (PFD) 153 compares the phase of a frequencyreference clock signal FREF 154 to the phase of a feedback clock signalFFB 155 and outputs up and down phase error pulse signals UP and DN.Change pump 156 and Low Pass Loop Filter (LPF) 157 convert the errorpulses into a relatively slowly changing error voltage. This errorvoltage is supplied onto the fine tuning input lead of VCO 111. VCO 111outputs a differential signal 158 of a frequency that corresponds to thelevel of the error voltage. Differential signal 158 includes two signalsof the same frequency FVCO but with relative phases of 0 degrees and 180degrees. A divide-by-two circuit 159 divides the differential signal 158by two and outputs a quadrature signal 160 of frequency FVCO/2. Thequadrature signal 160 includes four signals of the same frequency butwith relative phases of 0 degrees, 90 degrees, 180 degrees, and 270degrees. The zero degree phase signal is divided by prescaler 161, andthen is further divided down by loop divider 162 to generate thefeedback clock signal FFB 155. How loop divider 162 divides iscontrolled by a sigma delta modulator 163. The PLL is tuned by changingthe multi-bit digital control value loaded into DTOP block 243 viaconductors 150 and serial bus 148. The DTOP block 243 supplies a finetuning digital control value to the sigma delta modulator 163 viaconductors 232 and supplies a coarse tuning digital control value to theVCO 111 via conductors 233. Buffer 165 may be considered an outputbuffer portion of divide-by-two circuit 159. The triangle buffer symbols166-169 are high-level symbols. For more detail on the actual circuitryrepresented by these high-level symbols 165-169, see FIG. 6 and thefollowing text. Blocks 170 and 171 represent divide-by-two circuits.Each of these divide-by-two circuits receives a differential signal of agiven frequency and outputs quadrature signals of half that frequency.Quadrature signals LO1, LO2, LO3 and LO4 are supplied to transmitter104, transmitter 105, receiver 106 and receiver 107, respectively. Acrystal clock signal TCXO from a crystal clock signal source XTAL/SRC234 (such as a crystal oscillator) is received via conductor 240 and issupplied to a frequency doubler 241. Whether doubler 241 doubles thefrequency of the crystal clock signal or not is determined by digitalcontrol information stored in register 242 as determined by controlinformation received via serial bus 148 and the DTOP logic block 243.

FIG. 6 is a diagram that shows the LO signal generation and distributioncircuitry in more detail. VCO 111 is controlled to generate and outputthe differential signal 158 at about 10 GHz. This 10 GHz VCO frequencyis four times the desired local oscillator signal frequency used by thetransmitter or receiver if the transceiver is to be operating in thefirst frequency band (IEEE802.11b/g) and is two times the desired localoscillator signal frequency used by the transmitter or receiver if thetransceiver is to be operating in the second frequency band(IEEE802.11a).

Unlike the first architecture described above in the background section,this 10 GHz VCO signal is not routed large distances across theintegrated circuit. At a location very close to VCO 111, thedivide-by-two circuit 159 receives the differential signal VCO 158 anddivides it down in frequency by two, and outputs quadrature signal 160of half the frequency of signal 158. Quadrature signal 160 comprisesfour component signals of different phases (0, 90, 180, 270 degrees) butof the same frequency. Two of the four component signals of theresulting quadrature signal 160 can be considered together to be adifferential signal. Two of the four component signals are routed as adifferential signal 172 via buffer 173 and phase correcting circuitry174-177 to the divide-by-two circuit 170 located close to mixer 115 oftransmitter 104. The divide-by-two circuit 170 receives the phasecorrected version 179 of the differential signal, divides it down infrequency by two, and generates the quadrature signal 121 of a frequencyof half the frequency of signal 179.

Similarly, two others of the four component signals of quadrature signal160 can be considered together to be a differential signal. These twosignals are routed as a differential signal 181 via buffer 182 and phasecorrecting circuitry 183-186 to the divide-by-two circuit 171 locatedclose to mixer 134 of transmitter 106. The divide-by-two circuit 171receives the phase corrected version 188 of the differential signal,divides it down in frequency by two, and generates the quadrature signal138 of a frequency of half the frequency of signal 188. To cover thefirst band (IEEE802.11b/g) that extends from 2.412 GHz to 2.484 GHz, theVCO 111 is tunable to output VCO output signal 158 in the range of from9.648 GHz to 9.936 GHz.

For the transmitter and receiver of the second band (IEEE802.11a), the5.0 GHz quadrature local oscillator signal 160 is routed from the outputof divide-by-two circuit 159 to the mixers 125 and 142. Because therelative phases of the four component signals making up this quadraturesignal 160 can be disturbed or changed when the quadrature signal isrouted over such a long distance, phase mismatch correction circuitry isemployed to correct the relative phases of the component signalsrelative to one another so that the phases are correct when thequadrature signal is supplied to the mixers 125 and 142. For example,the phase mismatch corrected quadrature signal 130 supplied to mixer 125of transmitter 105 is communicated from divider 159 via two phasemismatch correction circuits. Two of the component signals making upquadrature signal 160 pass through a buffer 189 and phase mismatchcorrection circuitry 190-193 before being supplied to mixer 125. Twoothers of the component signals making up quadrature signal 160 passthrough a buffer 194 and phase mismatch correction circuitry 195-198before being supplied to mixer 125. The phase mismatch correctedquadrature signal 146 supplied to mixer 142 of receiver 107 iscommunicated from divider 159 via two other phase mismatch correctioncircuits. The two of the component signals making up quadrature signal160 that pass through buffer 189 are made to pass through phase mismatchcorrection circuitry 199-202 before being supplied to mixer 142. The twoof the component signals making up quadrature signal 160 that passthrough buffer 194 are made to pass through phase mismatch correctioncircuitry 203-206 before being supplied to mixer 142. The dark dashedvertical line 207 in FIG. 6 represents a relatively long distance ofrouting of more than one hundred microns. Divider 159, by contrast, islocated relatively close to (less than fifty microns from) VCO 111.Similarly, divider 170 is located relatively close to (less than fiftymicrons from) mixer 115 and divider 171 is located relatively close to(less than fifty microns from) mixer 134.

Although VCO 111 is operating at a high frequency (approximately 10 GHz)that is an integer multiple of the PA output signal frequency(approximately 5.0 GHz or approximately 2.5 GHz), the integer multipleis an even integer such as two or four. When the harmonic content of aPA output signal is analyzed, the odd order harmonic signal content isgenerally relatively stronger, whereas the even order harmonic signalcontent is generally relatively weaker. Although there is a fourth orderharmonic of the 2.5 GHz PA output signal at 10 GHz that would affect aVCO tuned and operating at 10 GHz, and although there is a second orderharmonic of the 5.0 GHz PA output signal at 10 GHz that would affectwith a VCO tuned and operating at 10 GHz, the fact that these harmonicsare even order harmonics means that they are of relatively weak signalstrength and consequently tend not to affect VCO operation to a largedegree. Stronger odd order harmonics in the PA output signal arestronger, but these odd order harmonics are not at 10 GHz so couplinginto the VCO that is tuned to operate at 10 GHz is relatively small. Theeffect of even order harmonics in the PA output signal is still furtherreduced due to the differential signal routing employed. The VCO isdesigned such that coupling gain of signals above 10 GHz back into thetank of the VCO drops off quickly for frequencies above 10 GHz. Thishelps further reduce the impact of any PA output signal harmonics thathave frequencies greater than 10 GHz (for example, a fifth orderharmonic of a 2.5 GHz PA output signal fundamental or a third orderharmonic of a 5.0 GHz PA output signal fundamental) on 10 GHz VCOoperation.

FIG. 7 is a simplified diagram of the LO generation and distributioncircuitry of FIG. 6. The VCO divide-by-two circuit 159 is located closeto the VCO 111 to generate a quadrature signal of frequency FVCO/2 fromthe VCO differential output signal 158 of frequency FVCO. For thetransmitter and receiver of the lower frequency band (the first band),differential signals of a higher frequency than is needed (FVCO/2) arerouted from the VCO's divider 159 to the transmitter and receiver, andthen divide-by-two circuits (170, 171) are used locally to generate theneeded quadrature signals of frequency FVCO/4 from the differentialsignals. The differential signals of frequency FVCO/2 may be routed fromVCO divider 159 to the local divide-by-two circuits 170 and 171 viabuffers, line drivers, and/or phase-mismatch correction circuitry.

For the transmitter and receiver of the higher frequency band (thesecond band), quadrature signals of the desired frequency (FVCO/2) arerouted as quadrature signals from the VCO divide-by-two circuit 159 tothe transmitter and receiver without local divide-by-two circuitry.Phase mismatch correction circuitry (190-193, 195-198, 199-202, 203-206)in the signal path of the quadrature signals corrects any phase mismatchthat may be introduced in the component signals of the quadraturesignals due to the long distance of routing. The quadrature signals offrequency FVCO/2 may be routed from VCO divider 159 to the transmitter105 and receiver 107 via buffers, line drivers and/or the phase-mismatchcorrection circuitry.

FIG. 8 is a diagram of one instance of phase mismatch correctioncircuitry 195-198. There are five other instances of this circuitry inthe distribution circuitry of FIG. 7. The phase mismatch correctioncircuitry 195-198 includes a first Programmable Driver (PD) 195, aTransmission Line (TL) 196, a Programmable Delay Line (PDL) 197, and asecond Programmable Driver (PD) 198. Transmission line 196 includes twolengths of metal conductors 238 and 239. Each of these metal conductorsis a single contiguous length of metal that has a controlled andsubstantially uniform impedance.

Programmable Delay Line 197 includes two sets of inverters. The firstset of these inverters is represented by the inverter symbol 208. Thesecond set of these inverters is represented by the inverter symbol 209.Each of these sets of inverters is a set of parallel-connected CMOSinverters, where individual ones of the inverters can be enabled ordisabled by digital control signals B[0-4] and BB[0-4]. One of thesesets of the inverters 210-229 is illustrated in the bottom portion ofFIG. 8. By adjusting the relative drive strength of the programmabledelay lines (PDL) in the signal propagation delay paths of the componentsignals of a quadrature signal, the relative phases of the componentsignals are adjusted so that the phases are at their correct 0, 90, 180and 270 degrees values at the location where the quadrature signal is toenter a mixer.

In one example, an internal loop back test connection 235 is used sothat receiver 107 can be used to receive a signal output by transmitter105. The signal is received through the RF transceiver integratedcircuit 102, and after downconversion and filtering is supplied viaconductors 144 to ADC 145 in the digital baseband integrated circuit103. ADC 145 digitizes the signal and the signal is demodulated. Phasemismatched detection and correction software 236 and processing thendetermines whether there is unwanted content in the demodulated output.If unwanted content is detected and this unwanted content is determinedto be due to I/Q phase mismatch, then the phase mismatch detection andcorrection software causes the phase mismatch correction circuitry195-198 and 190-193 supplying the I and Q signals to the transmitter 105change the relative phases between I and Q. The phase mismatch detectionand correction mechanism again uses receiver 107 to monitor thetransmission. The I and Q phases are adjusted until the unwanted contentin the demodulated signal is no longer present. For additional detailson a suitable phase mismatch detection and correction process, see:Behzad Razavi, “Design Consideration for Direct-Conversion Receivers”,IEEE Transactions On Circuits and Systems—II: Analog and Digital SignalProcessing, Vol. 44, No. 6, pages 428-435 (June 1997). I/Q mismatches inboth the signal LO4 supplied to receiver 107 as well as in signal LO2supplied to the transmitter 105 are corrected in this way. Referencenumeral 237 identifies a second internal loop back test connectionusable to perform I/Q mismatch correction on local oscillator signalsLO3 and LO1.

FIG. 9 is a diagram of one of the divide-by-two circuits, divider 159.Divide-by-two circuits 170 and 171 are of identical construction.Divide-by-two circuit 159 receives a differential signal on input leads301 and 302. Divide-by-two circuit 159 outputs a quadrature signal ofhalf the frequency of the input signal on output leads 303-306.

FIG. 10 is a more detailed diagram of the divide-by-two circuit 159. Thedivide-by-two circuit 159 includes two latches 307 and 308 coupledtogether as illustrated. As the input clock signal VOP/VON transitions,the first latch 307 is made to operate in the tracking mode and thesecond latch 308 is made to operate in the locking mode, and then whenthe input clock signal VOP/VON transitions again the first latch 307 ismade to operate in the locking mode and the second latch 308 is made tooperate in the tracking mode. In this way, the pair of latches operatesas a toggle flip-flop and frequency divides by two.

FIG. 11 is a more detailed diagram of one of the latches 307. The latch307 has two data input leads 309 and 310, two data output leads 313 and314, and two clock input leads 311 and 312. The latch includes a firstinverter including transistors 315 and 316, a second inverter includingtransistors 317 and 318, a first transmission gate including transistors319 and 320, a second transmission gate including transistors 321 and322, a third transmission gate including transistors 323 and 324, andfourth transmission gate including transistors 325 and 326. Depending onthe state of the differential input signal, the transparent latch 307either operates in a tracking mode or in a locking mode.

FIG. 12 is a diagram of operation of latch 307 in the tracking mode.Arrow 327 shows the signal path from input lead 309 to the input of theinverter 315, 316. Arrow 328 shows the signal path from input lead 310to the input of the inverter 317, 318. The differential output signaltracks the differential input signal.

FIG. 13 is a diagram of operation of latch 307 in the locking mode.Arrows 329 and 330 illustrate the signal path from the output of thefirst inverter 315, 316 to the input of the second inverter 317, 318,and from the output of the second inverter 317, 318 to the input of thefirst inverter 315, 316. The latch is locked due to the cross-couplingof the inverters.

FIG. 14 is a high-level diagram of one of the programmable drivers,programmable driver 195. Programmable driver 195 includes twodigitally-controllable inverting circuits 400 and 401 connected asillustrated.

FIG. 15 is a diagram of one of the two digitally-controllable invertingcircuits, circuit 400.

FIG. 16 is a diagram of the digitally-controllable inverting circuit 400of FIG. 15. The circuit includes a chain of three inverters 402-404between the input lead 405 and the output lead 406. In addition, thecircuit includes a second set of three inverters 407-409 that areconnected as a chain between input lead 405 and output lead 406, butthis second set of inverters can be enabled or disabled via transistors410 and 411 to increase or decrease the drive strength of the overallcircuit. To enable this second set of three inverters, digital signalSB[0] is made to be a digital logic low and digital signal S[0] is madeto be a digital logic high. In addition, the circuit includes a thirdset of two inverters 412 and 413 that are connected as a non-invertingchain between node 414 at the input of inverter 403 and the output lead406. To enable this third set of two inverters, digital signal SB[1] ismade to be a digital logic low to turn on transistor 415 and digitalsignal S[1] is made to be a digital logic high to enable transistor 416.Enabling the third set of two inverters increases the drive strength ofthe overall circuit.

FIG. 17 is a table that compares the amount of integrated circuit arearequired to implement the offset LO architecture of FIG. 1, theheterodyne LO architecture of FIG. 2, and the embodiment of FIG. 6.

FIG. 18 is a table that compares current consumption for the offset LOarchitecture of FIG. 1, the heterodyne LO architecture of FIG. 2, andthe embodiment of FIG. 6.

FIG. 19 is a flowchart of a method 500 of generating and distributinglocal oscillator signals in accordance with one novel aspect. In a firststep (step 501), a VCO is used to generate a first differential signalof frequency FVCO. In one example, the first differential signal issignal 158 of FIG. 6.

In a second step (step 502), the first differential signal is suppliedto a first divide-by-two circuit local to the VCO such that the firstdivide-by-two circuit outputs a first quadrature signal of frequencyFVCO/2. In one example, the first quadrature signal of frequency FVCO/2is signal 160 of FIG. 6.

In a third step (step 503), two of the four component signals of thefirst quadrature signal are supplied to a second divide-by-two circuitlocal to a first mixer of a first transmitter such that the seconddivide-by-two circuit outputs a second quadrature signal of frequencyFVCO/4 to the first mixer. In one example, the second divide-by-twocircuit is divider 170 of FIG. 6 and the first transmitter istransmitter 104 of FIG. 6.

In a fourth step (step 504), the first quadrature signal is supplied toa first phase mismatch correction circuit such that the first phasemismatch correction circuit outputs a first phase-corrected version ofthe first quadrature signal to a second mixer of a second transmitter.In one example, the first phase mismatch correction circuit iscorrection circuitry 195-198 and 190-193 and the first phase-correctedversion of the first quadrature signal is signal 130 of FIG. 6. In thisexample, the second transmitter is transmitter 105.

In a fifth step (step 505), two of the four component signals of thefirst quadrature signal are supplied to a third divide-by-two circuitlocal to a third mixer of a first receiver such that the thirddivide-by-two circuit outputs a third quadrature signal of frequencyFVCO/4 to the third mixer. In one example, the third divide-by-twocircuit is divider 117 of FIG. 6. In this example, the first receiver isreceiver 106.

In a sixth step (step 506), the first quadrature signal is supplied to asecond phase mismatch correction circuit such that the second phasemismatch correction circuit outputs a second phase corrected version ofthe first quadrature signal to a fourth mixer of a second receiver. Inone example, the second phase mismatch correction circuit is circuitry199-202 and 203-206. In this example, the second receiver is receiver107. The first transmitter, second transmitter, first receiver, andsecond receiver are parts of a multi-band 802.11 transceiver.

In one or more exemplary embodiments, the functions described may beimplemented in hardware, software, firmware, or any combination thereof.If implemented in software, the functions may be stored on ortransmitted over as one or more instructions or code on acomputer-readable medium. Computer-readable media includes both computerstorage media and communication media including any medium thatfacilitates transfer of a computer program from one place to another. Astorage media may be any available media that can be accessed by acomputer. By way of example, and not limitation, such computer-readablemedia can comprise RAM, ROM, EEPROM, CD-ROM or other optical diskstorage, magnetic disk storage or other magnetic storage devices, or anyother medium that can be used to carry or store desired program code inthe form of instructions or data structures and that can be accessed bya computer. Also, any connection is properly termed a computer-readablemedium. For example, if the software is transmitted from a website,server, or other remote source using a coaxial cable, fiber optic cable,twisted pair, digital subscriber line (DSL), or wireless technologiessuch as infrared, radio, and microwave, then the coaxial cable, fiberoptic cable, twisted pair, DSL, or wireless technologies such asinfrared, radio, and microwave are included in the definition of medium.Disk and disc, as used herein, includes compact disc (CD), laser disc,optical disc, digital versatile disc (DVD), floppy disk and blu-ray discwhere disks usually reproduce data magnetically, while discs reproducedata optically with lasers. Combinations of the above should also beincluded within the scope of computer-readable media.

In one example, memory 231 of FIG. 3 is a processor-readable medium thatstores a set of processor-executable instructions. When processor 151executes this set of instructions, the processor 151 is made to controlthe RF transceiver integrated circuit 102 via serial bus 148 suchthat: 1) in a first mode two of the four component signals of quadraturesignal 160 of frequency FVCO/2 are communicated a distance of more thanone hundred microns from divide-by-two circuit 159 to divide-by-twocircuit 170 such that divide-by-two circuit 170 supplies the quadraturesignal 121 of frequency FVCO/4 to mixer 115, and 2) in a second mode allfour of the component signals of quadrature signal 160 of frequencyFVCO/2 are communicated a distance of more than one hundred microns fromdivide-by-two circuit 159 through phase mismatch correction circuitry190-193, 195-198 to mixer 125. The processor 151 controls the drivers inthe signal paths such that signals are only supplied to the mixer of thetransmitter that is being used. Generally either transmitter 104 isbeing used, or transmitter 105 is being used, but both transmitters 104and 105 are not used simultaneously. In addition, the processor executesinstructions that cause the processor to detect I/Q phase mismatchconditions using test loop back connection 235 and receiver 107 and tocontrol the phase mismatch correction circuitry to eliminate or reduceI/Q phase mismatch in the four components of the quadrature signal 130as supplied to mixer 125.

Although certain specific embodiments are described above forinstructional purposes, the teachings of this patent document havegeneral applicability and are not limited to the specific embodimentsdescribed above. Although the LO generation and distributionarchitecture described above has special applicability to multi-bandIEEE802.11 transceivers, it is not limited thereto and has generalapplicability. Accordingly, various modifications, adaptations, andcombinations of the various features of the described specificembodiments can be practiced without departing from the scope of theclaims that are set forth below.

What is claimed is:
 1. An apparatus comprising: a Voltage ControlledOscillator (VCO) that outputs a differential VCO output signal offrequency FVCO; a first divide-by-two circuit coupled to receive thedifferential VCO output signal and to output a quadrature signal offrequency FVCO/2, wherein the quadrature signal comprises four componentsignals each having a frequency of FVCO/2; a first direct conversiondevice including a first mixer; and a second divide-by-two circuitcoupled to receive two of the four component signals of the quadraturesignal of frequency FVCO/2, and wherein the second divide-by-two circuitis coupled to supply a quadrature signal of frequency FVCO/4 to thefirst mixer.
 2. The apparatus of claim 1, wherein the first directconversion device is located more than one hundred microns away from thefirst divide by two circuit.
 3. The apparatus of claim 1, wherein thefirst divide-by-two circuit is located less than fifty microns away fromthe first direct conversion device.
 4. The apparatus of claim 1, furthercomprising: a second device including a second mixer.
 5. The apparatusof claim 4, further comprising: a first phase mismatch correctioncircuit coupled to receive the quadrature signal of frequency FVCO/2,wherein the first phase mismatch correction circuit is coupled to supplya first phase-corrected version of the quadrature signal of frequencyFVCO/2 to the second mixer.
 6. The apparatus of claim 4, wherein thesecond device is a direct conversion device.
 7. The apparatus of claim4, wherein the second device is a transmitter adapted to transmit in aband located at about FVCO/2.
 8. The apparatus of claim 1, furthercomprising: a third direct conversion device including a third mixer;and a third divide-by-two circuit coupled to receive two of the fourcomponent signals of the quadrature signal of frequency FVCO/2, andwherein the third divide-by-two circuit is coupled to supply aquadrature signal of frequency FVCO/4 to the third mixer.
 9. Theapparatus of claim 8, wherein the third direct conversion device is areceiver adapted to receive in a band located at about FVCO/4.
 10. Theapparatus of claim 8, further comprising: a first internal loopback testconnection usable in a phase correction process to supply a signaloutput by the first direct conversion device to third direct conversionreceiver.
 11. The apparatus of claim 8, further comprising: a fourthdirect conversion device including a fourth mixer; and a fourthdivide-by-two circuit coupled to receive two of the four componentsignals of the quadrature signal of frequency FVCO/2, and wherein thefourth divide-by-two circuit is coupled to supply a quadrature signal offrequency FVCO/4 to the fourth mixer.
 12. The apparatus of claim 11,wherein the fourth direct conversion device is a receiver adapted toreceive in a band located at about FVCO/2.
 13. The apparatus of claim11, further comprising: a second phase mismatch correction circuitcoupled to receive the quadrature signal of frequency FVCO/2, whereinthe second phase mismatch correction circuit is coupled to supply asecond phase-corrected version of the quadrature signal of frequencyFVCO/2 to the fourth mixer.
 14. The apparatus of claim 1, wherein thefirst direct conversion device is a first transmitter adapted totransmit in a band located at about FVCO/4.
 15. A method comprising:using a Voltage Controlled Oscillator (VCO) to generate a differentialVCO output signal of frequency FVCO; supplying the differential VCOoutput signal to a first divide-by-two circuit to obtain a quadraturesignal of frequency FVCO/2, wherein the quadrature signal comprises fourcomponent signals each having a frequency of FVCO/2; supplying two ofthe four signals of the quadrature signal to a second divide-by-twocircuit to obtain a quadrature signal of frequency FVCO/4; and supplyingthe quadrature signal of frequency FVCO/4 to a first direct conversiondevice including a first mixer.
 16. The method of claim 15, wherein thefirst direct conversion device is located more than one hundred micronsaway from the first divide: by: two circuit.
 17. The method of claim 15,wherein the first divide-by-two circuit is located less than fiftymicrons away from the VCO.
 18. The method of claim 15, furthercomprising: supplying the quadrature signal of frequency FVCO/2 to afirst phase mismatch correction circuit to obtain a firstphase-corrected version of the quadrature signal of frequency FVCO/2;and supplying the first phase-corrected version of the quadrature signalof frequency FVCO/2 to a second device including a second mixer.
 19. Themethod of claim 18, wherein the second device is a direct conversiondevice.
 20. The method of claim 18, wherein the second device is atransmitter adapted to transmit in a band located at about FVCO/2. 21.The method of claim 15, wherein the first direct conversion device is afirst transmitter adapted to transmit in a band located at about FVCO/4.22. An apparatus comprising: means for generating a differential VoltageControlled Oscillator (VCO) output signal of frequency FVCO; means fordividing the differential VCO output signal by two to obtain aquadrature signal of frequency FVCO/2, wherein the quadrature signalcomprises four component signals each having a frequency of FVCO/2; andmeans for dividing two of the four signals of the quadrature signal bytwo to obtain a quadrature signal of frequency FVCO/4, wherein the meansfor dividing two of the four signals is configured to supply thequadrature signal of frequency FVCO/4 to a first direct conversiondevice including a first mixer.
 23. The apparatus of claim 22, whereinthe first direct conversion device is located more than one hundredmicrons away from the means for dividing the differential VCO outputsignal by two.
 24. The apparatus of claim 22, wherein the means fordividing the differential VCO output signal by two is located less thanfifty microns away from the means for generating a differential VoltageControlled Oscillator (VCO) output signal.
 25. The apparatus of claim22, wherein: the means for dividing the differential VCO is configuredto supply the quadrature signal of frequency FVCO/2 to a phase mismatchcorrection circuit to obtain a phase-corrected version of the quadraturesignal of frequency FVCO/2; and the phase mismatch correction circuit isconfigured to supply the phase-corrected version of the quadraturesignal of frequency FVCO/2 to a second direct conversion deviceincluding a second mixer.
 26. The apparatus of claim 25, wherein thesecond direct conversion device is a transmitter adapted to transmit ina band located at about FVCO/2.
 27. The apparatus of claim 22, whereinthe first direct conversion device is a first transmitter adapted totransmit in a band located at about FVCO/4.